Method and device for transmission with reduced crosstalk

ABSTRACT

The invention relates to a method and a device for transmission with reduced crosstalk in interconnections used for sending a plurality of signals, such as the interconnections made with flat multiconductor cables, or with the tracks of a printed circuit board, or inside an integrated circuit. An interconnection with four parallel transmission conductors plus a reference conductor has each of its ends connected to a termination circuit. The transmitting circuit receives at its input the signals of the four channels of the source and its output terminals are connected to the conductors of the interconnection. The receiving circuit&#39;s input terminals are connected to the conductors of the interconnection, and its four output channels are connected to the destination. The signals of the four channels of the source are sent to the four channels of the destination, without noticeable crosstalk.

FIELD OF THE INVENTION

The invention relates to a method and a device for transmission withreduced crosstalk through interconnections used for sending a pluralityof signals, such as the ones made with flat multiconductor cables, orwith the tracks of a printed circuit board, or inside an integratedcircuit.

The French patent application number 0300064 of 6 Jan. 2003, entitled“Procédé et dispositif pour la transmission avec une faible diaphonie”is incorporated by reference.

PRIOR ART

Let us first consider the theoretical problem of an interconnection withn transmission conductors placed close to a reference conductor. Let usnumber these conductors from 0 to n, where 0 is the “referenceconductor” which will be used as a reference for measuring voltages, andwhich is often called the ground conductor.

As an example, we have represented in FIG. 1 an interconnection withfour parallel transmission conductors (1) between a source (2) and adestination (3). For instance, the source (2) may be composed of thefour output circuits of an integrated circuit, the destination (3) maybe composed of the four input circuits of an other integrated circuit,and the transmission conductors numbered 1, 2, 3 and 4 (this numberingdoes not show up in FIG. 1) may be the traces of a printed circuit boardon which the two integrated circuits are soldered, the conductor 0 beinga ground plane of this printed circuit board. Such an interconnectionmay clearly convey analog or digital signals. It is well known that,when the maximum frequency of the spectrum of the signals to be sentcorresponds to a wavelength that is not very big compared to the lengthof the interconnection, it is useful to provide for the implementationof terminations (4) at the ends of the interconnection, theseterminations being for instance made of resistors placed between each ofthe transmission conductors numbered from 1 to 4 and the referenceconductor.

Note that such terminations are sometimes included in the circuits ofthe source and/or in the circuits of the destination. Note also that insome cases, a single termination is used.

As shown in the example in FIG. 2, an interconnection (1) may also beconnected for instance to a plurality of line transmitters (2) and linereceivers (3), where the line transmitters and/or line receivers arespread over the length of the interconnection. Architectures referred toas “data bus” are of this type. The techniques that make this type ofstructure possible, for instance the techniques through which the outputof certain digital circuits can have a “high impedance” state, are wellknown. In the example in FIG. 2, the interconnection is terminated witha termination (4) at each end, as the one given in the example in FIG.1.

We define any point along an interconnection of length L with a realcurvilinear abscissa z, the interconnection extending from z=0 to z=L.

Any integer j greater than or equal to 1 and less than or equal to ncorresponds to the number of a transmission conductor of theinterconnection, that is to say to a conductor other than the referenceconductor. This integer may therefore be used as an index in order todefine, for each transmission conductor, two electrical variables, i.e.one current and one voltage. At a given abscissa z along the cable, wedefine in this manner the current i_(j) flowing in the transmissionconductor, and the voltage v_(j) between the transmission conductor andthe reference conductor. These n currents and these n voltages arerespectively called natural currents and natural voltages. The wording“natural electrical variable” will indiscriminately designate a naturalcurrent or a natural voltage.

In order to clarify our vocabulary, we will now present some basis ofthe matrix theory of multiconductor transmission lines, which is wellknown to specialists. Elements of this theory are for instance presentedin the book Analysis of Multiconductor Transmission Lines of C. R. Paul,published by John Wiley & Sons in 1994. When an interconnection canapproximately be considered as having characteristics that are uniformover its length (that is to say independent of z), its characterizationfor the transmission of signals and for crosstalk may be obtained with aper-unit-length inductance matrix L, a per-unit-length resistance matrixR, a per-unit-length capacitance matrix C, and a per-unit-lengthconductance matrix G, all being independent of z. Specialists refer inthis case to a uniform multiconductor transmission line. These matricesare symmetrical square matrices of order n, and they arefrequency-dependent. The matrices L, R, C and G may be used to write twoequations containing the column-vector I of the natural currents i₁. . ., i_(n) and the column-vector V of the natural voltages v₁, . . . ,v_(n) considered at the same abscissa z. We will therefore qualify thesefour matrices as “natural”. These two equations are called telegrapher'sequations by specialists, and may be written: $\begin{matrix}\{ \begin{matrix}{\frac{\mathbb{d}V}{\mathbb{d}z} = {{- ( {R + {j\quad\omega\quad L}} )}I}} \\{\frac{\mathbb{d}I}{\mathbb{d}z} = {{- ( {G + {j\quad\omega\quad C}} )}V}}\end{matrix}  & (1)\end{matrix}$where ω is the radian frequency.

We shall now use Z=R+jωL to denote the per-unit-length impedance matrixand Y=G+jωC to denote the per-unit length admittance matrix. It is wellknown to specialists that the equation (1) may be solved easily using asuitable diagonalization of the matrices ZY and YZ. The eigenvectorsobtained in this manner define the propagation modes, and theeigenvalues correspond to the propagation constants. More precisely, weshall use T and S to denote two regular matrices such that:$\begin{matrix}\{ {\begin{matrix}{{T^{- 1}{YZT}} = D} \\{{S^{- 1}{ZYS}} = D}\end{matrix}\quad{where}}  & (2) \\{D = {{diag}_{n}( {\gamma_{1}^{2},\ldots\quad,\gamma_{n}^{2}} )}} & (3)\end{matrix}$is the diagonal matrix of order n of the eigenvalues. These eigenvaluesare the squares of the propagation constants γ_(j) for specific waves,which we shall identify later, propagating toward the far end (that isto say toward z=L). The matrices Z and Y being symmetrical, we observethat if we determine, with a diagonalization of the matrix YZ, a matrixT satisfying the first line of the equation (2), thenS=^(t)T⁻¹   (4)is one solution of the second line of the equation (2). This shows thatif YZ is diagonalizable (which has been shown in interesting cases bymany authors), then YZ and ZY are diagonalizable into the same matrix D.The use of equation (4) is not in any way necessary for solving theequation (2), and other choices are possible. Thus, for instance,another possible choice for obtaining a solution S for the second lineof the equation (2) from a solution T of its first line isS=jωc_(KY) ⁻¹T   (5)where c_(K) is an arbitrary scalar different from zero, which may dependon the frequency, and which has the dimensions of a per-unit-lengthcapacitance.

The matrices T and S solutions of the equations (2) and (3) define a“modal transform” for the natural currents and for the natural voltages,and the results of this transform are called modal currents and modalvoltages. If we use I_(M) to denote the vector of the n modal currentsi_(m1), . . . , i_(Mn) and V_(M) to denote the vector of the n modalvoltages v_(M1), . . . , v_(Mn), we get: $\begin{matrix}\{ \begin{matrix}{V = {SV}_{M}} \\{I = {TI}_{M}}\end{matrix}  & (6)\end{matrix}$

Consequently, we shall call S the “transition matrix from modal voltagesto natural voltages”, and T the “transition matrix from modal currentsto natural currents” (for comparison with the French patent applicationnumber 0300064 of 6 Jan. 2003, it is useful to note that the transitionmatrix from the basis C to the basis B is called “matrice de passage dela base B a 1a base C” in French). The modal currents and the modalvoltages have the remarkable property of being able to propagate alongthe transmission line without coupling to one another when they have adifferent index. We can point out that for a given j, a modal currenti_(Mj) and a modal voltage v_(Mj) propagate with the same propagationconstant γ_(j) toward the far end (toward z=L), and with the oppositepropagation constant −γ_(j) toward the near end (toward z=0). Thewording “modal electrical variable” will indiscriminately designate amodal current or a modal voltage. The matrices S and T are therefore thetransition matrices from modal electrical variables to naturalelectrical variables.

We shall note that the equation (2) means that the column-vectors of S(respectively, of T) are linearly independent eigenvectors of ZY(respectively, of YZ), and consequently S and T are not defined in aunique manner by the equations (2) and (3) only, because: first theorder of the eigenvalues in the equation (3) is arbitrary, and secondthe choice of eigenvectors corresponding to a degenerate eigenvalue isarbitrary. The implementation of an additional condition such asequation (4) or equation (5) does not remove this indetermination.

In order to indicate that a matrix S and a matrix T are defined by therelations (2), (3) and (5) we shall say that they are “associated”. Inthis case, it is clear that for any integer j between 1 and n the j-thcolumn-vector of S corresponds to the same eigenvalue as the j-thcolumn-vector of T.

As from the equations (1), (2) and (3), it is possible to define thecharacteristic impedance matrix Z_(c) of the multiconductor transmissionline, as:Z_(c)=SΓ⁻¹S⁻¹Z=SΓS⁻¹Y⁻¹=Y⁻¹TΓT⁻¹=ZTΓ⁻¹T⁻¹   (7)whereΓ=diag_(n)(γ₁, . . . , γ_(n))   (8)is the diagonal matrix of order n of the propagation constants γ_(i),which have the dimensions of the inverse of a length. Thischaracteristic impedance matrix is such that:

-   a) for any wave propagating on the multiconductor transmission line    toward increasing z, the column-vector of the natural voltages V⁻ is    related to the column-vector of the natural currents I⁻ by:    V⁺=Z_(C)I⁺  (9)-   b) for any wave propagating on the multiconductor transmission line    toward decreasing z, the column-vector of the natural voltages V⁻ is    related to the column-vector of the natural currents I⁻ by:    V ⁻ =−Z _(C)I⁻  (10)

Using a well-known reasoning, one obtains that at one end of themulticonductor transmission line connected to a linear (n+1)-terminaldevice (one terminal of which is connected to the reference conductor,and the n other terminals of which are connected to the n transmissionconductors) presenting to the multiconductor transmission line animpedance matrix equal to its characteristic impedance matrix, noreflection occurs for incident waves.

It is also possible to show that, when the matrix S and the matrix T areassociated:

-   c) for any wave propagating on the multiconductor transmission line    toward increasing z, the column-vector of the modal voltages V_(M) ⁺    is related to the column-vector of the modal currents I_(M) ⁺ by:    $\begin{matrix}    {V_{M}^{+} = {\frac{1}{j\quad\omega\quad c_{K}}\Gamma\quad I_{M}^{+}}} & (11)    \end{matrix}$-   d) for any wave propagating on the multiconductor transmission line    toward decreasing z, the column-vector of the modal voltages V_(M) ⁻    is related to the column-vector of the modal currents I_(M) ⁻ by:    $\begin{matrix}    {V_{M}^{-} = {{- \frac{1}{j\quad\omega\quad c_{K}}}\Gamma\quad I_{M}^{-}}} & (12)    \end{matrix}$

It is necessary to state that, according to the theory of multiconductortransmission lines, the presence of a reference conductor is necessary.However, a priori no specific physical characteristic distinguishes whatwe have called a transmission conductor (which some authors call asignal conductor) from the reference conductor. Designating a conductoras the reference conductor is only a theoretical requirement. Inpractice however, we note that electronic apparatuses often make aspecific use of the ground of a circuit, which is a set ofinterconnected conductors, because the circuits make a preferred use ofthe voltages defined with respect to the ground. Whenever possible, itis therefore natural to select the ground as the reference conductor. Weshall also note that a conductor of the interconnection other than thereference conductor is always called transmission conductor”, and thatthis does not mean that it is necessarily used for the transmission of asignal. It is for instance customary to connect some of the transmissionconductors to ground, in order to reduce the crosstalk

It is also important to clearly distinguish the interconnection, aphysical device implementing conductors and insulators, from the modelwhich describes some of its properties, in this case the model of themulticonductor transmission line uniform over its length. Besides, thismodel is not capable of describing all interconnections. One can showthat this model is well suited for describing the behavior ofinterconnections whose conductors are all parallel cylinders (notnecessarily of revolution) sufficiently close with respect to thewavelength of the signal being considered, these conductors beingsurrounded by dielectrics the characteristics of which are uniform overthe length of the interconnection. This model may also appropriatelydescribe interconnections made of conductors which are parallel andsufficiently close over only the greatest part of their length, and alsoother types of interconnection.

The person skilled in the art knows that it is generally necessary toinclude all conductors between which a significant coupling is likely tooccur, in the multiconductor transmission line model. Thus, according toa first example, an unshielded flat cable with 8 conductors laying flaton a flat conductor over all its length must normally be treated as aninterconnection having 9 conductors including the reference conductor,even if one of the conductors of the flat cable has been designated asreference conductor. According to a second example, if a secondunshielded flat cable with 8 conductors is flattened against the firstone, the whole must normally be treated as an interconnection with 17conductors. According to a third example, when a multiconductor cablehas a screen surrounding its internal conductors, this screen must beregarded as one of the conductors of the interconnection.

We note that the reference conductor is sometimes made of severalsufficiently interconnected conductors. This is for instance the casewith the stripline structure well known to the person skilled in theart, in which the reference conductor is made of two interconnectedground planes. By the same token, it is appropriate to treat as a singlereference conductor a plurality of conductors between which a lowimpedance is maintained in the operating frequency band, at a sufficientnumber of points along the direction of propagation. As an example, in amultilayer printed circuit board, the traces of an internal layer, usedas transmission conductors, may be routed between a conducting planeused for the ground (ground plane) and a conducting plane connected to apower supply voltage. The person skilled in the art knows that if a lowimpedance is maintained between these conducting planes by severaldecoupling capacitors connected between these conducting planes andspread over along internal traces, then the two conducting planes,though at different potentials, behave indeed as a single referenceconductor for the propagation of signals. In the following, the wording“a reference conductor” may therefore designate a reference conductorconnected to one or several other conductors, at a sufficient number ofpoints along the direction of propagation, through impedancessufficiently low in the frequency band of operation.

The elementary theoretical principles which we have just presented arethe basis of a computation method which enables predicting crosstalk ininterconnections. In the case of interconnections used for transmittinga plurality of signals, the crosstalk is an undesirable phenomenon, anddesigners try to minimize it, as far as possible. The state of the artas regards fighting against crosstalk in interconnections mainlyimplements the following techniques:

-   1) the use of balanced transmission lines, also called symmetrical    transmission lines, to which sources of differential signals and    receivers of differential signals are connected, as explained for    instance in Chapter 4 of the book Noise Reduction Techniques in    Electronic Systems by H. W. Ott, second edition, published by John    Wiley & Sons in 1988;-   2) the termination of each pair of a set of balanced transmission    lines by its connection to a “matched impedance” between the two    conductors of the pair, at one and/or the other end, as in the    termination of telephone lines (which also reduces the echo    problem);-   3) in the case of unbalanced transmission lines, by increasing the    distance between each of the transmission conductors 1 to n, for    instance by moving the traces corresponding to these transmission    conductors away from each other in the case of printed circuit    board;-   4) in the case of unbalanced transmission lines, by decreasing the    distance between each of the transmission conductors 1 to n and the    reference conductor, for instance by using as the reference    conductor a ground plane layer under the traces corresponding to the    transmission conductors 1 to n, in the case of a printed circuit    board;-   5) in the case of unbalanced transmission lines, by decreasing the    bandwidth used for the signals;-   6) in the case of unbalanced transmission lines, by terminating each    transmission conductor by a connection to a linear dipole with a    “matched impedance”, the other terminal of this linear dipole being    connected to the reference conductor (which also reduces the echo    problem);-   7) by using grounded conductors to separate the signals to be sent,    for instance in cables including conductors dedicated to the    function of screening, called screens, or according to an other    example, by using some of the transmission conductors as screens.

The techniques of 1) and 2) above, which are often implemented ontwisted pair cables, achieve excellent performance, but they require theuse of two transmission conductors for each signal to be sent, which iseconomically disadvantageous. They are difficult to implement beyond afew hundred megahertz. The technique of 7) is also expensive.

The techniques of 3) and 4) above are efficient when used jointly, butthis approach takes up room, a factor that is difficult to accept incurrent printed circuit boards as well as in cables. The technique of 5)cannot be used in many situations in which the characteristics of thesignals, and therefore their spectrum, are specified.

The techniques of 2) and of 6) above are based on a simple principle:the echo waves are undesirable and they themselves give rise to somecrosstalk. Decreasing the echo waves therefore enables reducing thecrosstalk. It is also necessary to clarify what is meant by “matchedimpedances” in the wording of these techniques: it is a question of theimpedance of a dipole which is used to minimize the reflections of asignal on the pair being considered in the case of technique 2), or onthe transmission conductor being considered in the case of technique 6).The authors who are accurate on this point generally consider that thevalue to be assigned to the “matched impedance” of a transmissionconductor is the characteristic impedance of the line with a singletransmission conductor (plus a return conductor e.g. the referenceconductor) obtained when the propagation on the other transmissionconductors of the interconnection is not considered. The person skilledin the art understands that this point of view is an approximation whichcan only be fully justified in the case where the coupling with theother transmission conductors is very weak. In general, these “matchedimpedances” do not produce a termination showing an impedance matrixnear the characteristic impedance matrix to the interconnection. Forinstance, the terminations (4) of the device presented in FIG. 1 andFIG. 2 are made of “matched impedances” of this type.

Note finally that according to variations of technique 6), designers, inorder to limit the power consumed by a signal present at the terminalsof such a “matched load”, may replace it with a resistance in serieswith a capacitor, in such a way that the matched impedance only appearsat the highest frequencies of the spectrum of the signals transmitted,frequencies for which the crosstalk is often the most detrimental. Otherdesigners also use non linear terminations, for instance using diodes.

One may say that these techniques are limited in the following manner:their performance levels are low, or they require a substantial crossdimension of the interconnection, because of the increased spacing ofthe transmission conductors, or because of the use of a much largernumber of conductors (typically twice the number) than the number ofsignals to be sent.

DESCRIPTION OF THE INVENTION

The purpose of the method of the invention is the transmission withreduced crosstalk through interconnections with two or more transmissionconductors, without the limitations of these known techniques.

The invention is about a method for transmitting through aninterconnection with n transmission conductors and a referenceconductor, n being an integer greater than or equal to 2, the methodproviding, in a known frequency band, m transmission channels eachcorresponding to a signal to be sent from the input of at least onetransmitting circuit to the output of at least one receiving circuit,where m is an integer greater than or equal to 2 and less than or equalto n, the method comprising the steps of:

-   -   modeling the interconnection, taking into account the lumped        impedances seen by the interconnection and caused by the        circuits connected to the interconnection elsewhere than at the        ends of the interconnection, as a multiconductor transmission        line having uniform electrical characteristics over its length        for the known frequency band;    -   determining, for the multiconductor transmission line and the        known frequency band, the characteristic impedance matrix and a        transition matrix from modal electrical variables to natural        electrical variables;    -   placing at at least one end of the interconnection, a        termination circuit having an impedance matrix approximating the        characteristic impedance matrix;    -   combining the m input signals in one of the transmitting        circuits without using a transformer for this purpose, according        to linear combinations defined by the transition matrix from        modal electrical variables to natural electrical variables, so        as to obtain at the output of said one of the transmitting        circuits, output being connected to the n transmission        conductors, the generation of modal electrical variables, each        being proportional to a single signal among the input signals;        and    -   combining in one of the receiving circuits, the input of which        is connected to the n transmission conductors, without using a        transformer for this purpose, the signals present on the        transmission conductors, according to linear combinations        defined by the inverse of the transition matrix from modal        electrical variables to natural electrical variables, so as to        obtain at the output of said one of the receiving circuits m        output signals each corresponding to one of the transmission        channels, each output signal being proportional to a single        modal electrical variable among the modal electrical variables.

The person skilled in the art fully understands the principlesimplemented by the invention. It uses a superposition of waves beingeach composed of a unique modal electric variable corresponding to achannel, because these waves have the following properties:

-   a) the wave of a modal electrical variable propagates along the    multiconductor transmission line without being coupled to other    modal electrical variables of a different index, which follows from    the explanations given above after the equation (6);-   b) at one end of the multiconductor transmission line connected to a    termination circuit presenting an impedance matrix near the    characteristic impedance matrix, the wave of a modal electrical    variable is absorbed, without giving rise to any significant    reflected wave, which results from the property stated above, after    the equations (9) and (10).

These properties show that the propagation of waves each correspondingto a single modal variable, produced with a suitable conversion in oneof the transmitting circuits and used with an inverse conversion in oneof the receiving circuits enables transmission without crosstalk betweenthe channels.

Any of the n natural voltages (respectively, natural currents) being alinear combination of the n modal voltages (respectively, modalcurrents), according to equation (6), it appears that the value of anatural electrical variable a priori depends on the value of each of thesignals present on each of the n channels. This is radically differentfrom the behavior expected from devices such as those presented in FIG.1 and FIG. 2.

The person skilled in the art understands that the function of thetermination circuits is to ensure that no reflection of an incidentsignal of a disturbing level occurs at an end of the interconnection. Itis clear that the lower the desired maximum crosstalk coupling level,the lower will be the level of reflection of incident signals which willhave to be regarded as disturbing, and that, in order not to exceed thislevel, it must be specified that the termination circuit must have animpedance matrix that is closer to the characteristic impedance matrix.

In order to ensure that no reflection of an incident signal of adisturbing level occurs at an end of the interconnection, the personskilled in the art understands that it is sufficient, when one or moretransmitting circuits are connected at a single end of theinterconnection, to arrange a termination circuit presenting animpedance matrix sufficiently close to the characteristic impedancematrix, at the other end of the interconnection. The person skilled inthe art also sees that in all other cases, that is to say when atransmitting circuit is connected elsewhere than at one end of theinterconnection, and/or when transmitting circuits are connected at bothends of the interconnection, it is necessary, in order to ensure that noreflection of an incident signal of a disturbing level occurs at an endof the interconnection, to arrange a termination circuit presenting animpedance matrix sufficiently close to the characteristic impedancematrix at both ends of the interconnection.

Thus, according to the method of the invention we may:

-   -   either arrange at one end only of the interconnection, a        termination circuit with an impedance matrix near the        characteristic impedance matrix    -   or arrange at both ends of the interconnection, a termination        circuit with an impedance matrix near the characteristic        impedance matrix.

In order that this principle results in the desired characteristics, itis important that the interconnection behaves like a multiconductortransmission line uniform over its length, because a lack of homogeneitysuch as a variation of the characteristic impedance matrix with respectto z may produce detrimental couplings between the channels, that is tosay, crosstalk. Note that the effect of inhomogeneity depends on thewavelength of the signals transmitted, and therefore on their frequency,as it is legitimate to consider that, in practice, the waves are onlyaffected by a moving average of the natural matrices along theinterconnection, over a distance corresponding to a fraction of thewavelength. Consequently, on a perfectly uniform interconnection exceptfor a localized inhomogeneity, the effect of the inhomogeneity isrelatively less at lower frequencies, given that, for biggerwavelengths, the effect of the inhomogeneity is smoothed by an averagingon a greater length. This phenomenon is beneficial because inhomogeneityis unavoidable in practice, for instance at the connection point of atransmitting circuit or of a receiving circuit. For instance, such ainhomogeneity may correspond to a lumped capacitance matrix caused bythe transmitting circuit or by the receiving circuit, corresponding tolumped impedances.

In some cases, in order to take into account the lumped impedances seenby the interconnection and caused by the circuits connected to itelsewhere than at its ends, the designer need only observe that they arenot present or that they may be ignored. In other cases, in order totake into account the lumped impedances seen by the interconnection andcaused by the circuits connected to it elsewhere than at its ends, thedesigner must quantitatively consider these lumped impedances to obtaina multiconductor transmission line having sufficiently uniformelectrical characteristics over its length. For instance, theinterconnection could see a receiving circuit as a capacitance matrixadding to its own capacitance matrix: this lumped capacitance couldtherefore be offset by a suitably proportioned local modification of thegeometrical characteristics of the interconnection in the vicinity ofthe connection point. As a second example, capacitance matriceslocalized at connection points regularly spaced along theinterconnection could be taken into account to obtain a prescribedaverage p.u.l. capacitance matrix relevant up to a given maximumfrequency, by using suitably proportioned transmission conductors.

According to the invention, the modal electrical variables generated bya transmitting circuit are each proportional to a single signal amongthe input signals. Since m signals must be sent, there are at least mmodal electrical variables. According to the method of the invention, itis possible, in particular, to obtain the generation of m modalelectrical variables at the output of a transmitting circuit. This maybe the most economical procedure. However, it is also conceivable, whenm is less than n, that more than m modal electrical variables aregenerated for the m input signals.

According to the method of the invention, the number m of transmissionchannels between any one of the transmitting circuits and any one of thereceiving circuits may be equal to the number n of transmissionconductors. This method is preferred because it is generally the mosteconomical. However, it is also conceivable to use a number n oftransmission conductors, greater than the number m of channels (thiscase is mentioned in the “Indications on Industrial Applications”section below).

According to the method of the invention, the electrical variables maybeeither all voltages or all electric currents. Note that, when thematrices S and T are associated, according to the equations (11) and(12), considering that Γ is a diagonal matrix, we may say that in agiven direction of propagation, for any integer j between 1 and n, themodal voltage v_(Mj) is proportional to the modal current i_(Mj).Therefore:

-   it is physically equivalent that a transmitting circuit “generates    modal voltages on the transmission conductors, where each modal    voltage is proportional to only one of the input signals”, or that    it “generates modal currents on the transmission conductors, where    each modal current is proportional to only one of the input    signals”, and-   it is physically equivalent that a receiving circuit delivers at its    output “m output signals each corresponding to one of the    transmission channels, where each output signal is proportional to    only one of the modal voltages”, or that it delivers at its output    “m output signals each corresponding to one of the transmission    channels, where each output signal is proportional to only one of    the modal currents”.

Therefore, the use of either currents or voltages as electricalvariables is without physical effect. From the standpoint of design, itcould be more pleasant to use currents or voltages depending on the typeof device selected to implement the method. For instance, to proportiona transmitting circuit presenting a low impedance to theinterconnection, the designer may prefer to speak of modal voltages,whereas, on the contrary, to proportion a transmitting circuitpresenting a high impedance to the interconnection, the designer mayprefer to speak of modal currents.

According to the method of the invention, conductors and dielectrics maybe used such that the section of the interconnection in a planeorthogonal to the direction of propagation does not change, except for ascale factor, over the greatest part of the length of theinterconnection, in the vicinity of the transmission conductors. Theperson skilled in the art knows that this condition indeed allowsmaintaining practically uniform electrical characteristics over thelength of the interconnection. This condition especially includes thestandard case of straight interconnection, therefore parallel to anaxis, and invariant by a translation along this axis, in the vicinity ofthe transmission conductors. But contrary to this case, this conditionalso includes the case where the transmission conductors are curved.Finally, note that this condition is generally not compatible with theimplementation of a balanced transmission line, because the balance ofthe interconnection, which implies notably that some p.u.l capacitancevalues be equal, depends upon the technique of twisting. There arehowever some exceptions where this condition can be met with a balancedinterconnection. For instance, for n=2, an exception of this typeimplements an interconnection made of two straight transmissionconductors parallel to a ground plane constituting the referenceconductor, where the interconnection is symmetrical with respect to aplane P orthogonal to the ground plane, the plane P containing astraight line parallel to the transmission conductors.

Detailed examples given below include a schematic diagram according tothe invention, in the case of a balanced interconnection for which n=2and in the case of an interconnection for which n=3. When n is greaterthan or equal to three, the interconnections are often not balanced. Itis therefore important to note that, according to the method of theinvention, n may be greater than or equal to three.

Note that, in many possible cases, as the person skilled in the artknows, we can consider that, when computing the matrices Z_(C), S and Tof the multiconductor transmission line, the losses are negligible insome frequency bands, for instance for frequencies greater than 100 kHz,and that in this case, the characteristic impedance matrix is real andfrequency-independent, and the matrices S and T chosen may be real andfrequency-independent. Conversely, at frequencies lower than 10 kHz,losses are often not negligible and the characteristic impedance matrixcannot be considered as real, which obviously leads to a more compleximplementation of the method of the invention. However, this questioncan often be disregarded, because the crosstalk at frequencies lowerthan 10 kHz may in many cases be ignored, and in these cases, it may beof no importance that the termination circuits arranged at one end orboth ends of the interconnection present an impedance matrix near thecharacteristic impedance matrix at these frequencies.

The method of the invention is therefore particularly suited to the casewhere the known frequency band contains frequencies ranging from 100 kHzto 100 GHz.

The person skilled in the art knows, for instance, from a computationbased on the geometry of the conductors and insulators, on theconductivity of the conductors and on the permittivity and the losses ofthe insulators, how to determine the natural matrices L, R, C and G of amulticonductor transmission line, as a function of frequency. The personskilled in the art also knows how to measure these matrices. It istherefore clear that it is possible to accurately determine thecharacteristic impedance matrix of the multiconductor transmission linein any frequency interval, up to the maximum frequency for which thetransmission line theory is applicable. This maximum frequency dependson the cross dimensions of the interconnection, and the person skilledin the art knows that it corresponds to the appearance of the firstnon-evanescent propagation modes other than the quasi-TEM modes. In thissame frequency interval, it is obviously also possible to determine a“transition matrix from modal voltages to natural voltages” S and/or a“transition matrix from modal currents to natural currents” T, as afunction of frequency, so as to define modal voltages and/or modalcurrents.

The characteristic impedance matrix and a suitable choice of thematrices S and/or T may therefore be determined, for instance, in twodistinct contexts: firstly when the interconnection has been chosen andthe method of the invention must be applied to the interconnection byadapting the other parts of a device implementing this method, secondlywhen the parts of a device implementing this method, other than theinterconnection, have been defined beforehand, and an appropriateinterconnection should be designed.

A device for proportioning the circuits used in a method of theinvention is described in the next sentence. A device for proportioningthe circuits used in a method for transmitting through aninterconnection with n transmission conductors and a referenceconductor, n being an integer greater than or equal to 2, the methodproviding, in a known frequency band, m transmission channels eachcorresponding to a signal to be sent from the input of at least onetransmitting circuit to the output of at least one receiving circuit,where m is an integer greater than or equal to 2 and less than or equalto n, may comprise:

-   -   means for modeling the interconnection, taking into account the        lumped impedances seen by the interconnection and caused by the        circuits connected to the interconnection elsewhere than at the        ends of the interconnection, as a multiconductor transmission        line having uniform electrical characteristics over its length        for the known frequency band;    -   means for determining, for the multiconductor transmission line        and the known frequency band, the characteristic impedance        matrix and a transition matrix from modal electrical variables        to natural electrical variables;    -   means for proportioning a termination circuit having an        impedance matrix approximating the characteristic impedance        matrix;    -   means for proportioning one of the transmitting circuits which        combines the m input signals, without using a transformer for        this purpose, according to linear combinations defined by the        transition matrix from modal electrical variables to natural        electrical variables, so as to obtain at the output of said one        of the transmitting circuits, output being connected to the n        transmission conductors, the generation of modal electrical        variables, each being proportional to a single signal among the        input signals; and    -   means for proportioning one of the receiving circuits, the input        of which is connected to the n transmission conductors, which        combines, without using a transformer for this purpose, the        signals present on the transmission conductors according to        linear combinations defined by the inverse of the transition        matrix from modal electrical variables to natural electrical        variables, so as to obtain at the output of said one of the        receiving circuits m output signals each corresponding to one of        the transmission channels, each output signal being proportional        to a single modal electrical variable among the modal electrical        variables.

The device for proportioning the circuits used in a method of theinvention may be such that the means for modeling the interconnectioncomprise means for measuring and/or for computing the real electricalcharacteristics of the interconnection, based on the relative layout ofthe transmission conductors and the reference conductor, and on thecharacteristics of the dielectrics surrounding them.

The device for proportioning the circuits used in a method of theinvention may be such that the means for modeling the interconnectioncomprise:

-   -   means for calculating one or more error coefficients for        variance between the actual electrical characteristics of the        interconnection and the desired characteristics, for the known        frequency band; and    -   means for optimizing the relative position of the transmission        conductors by minimizing the error coefficients or coefficients.

A device for implementing the method of the invention is described inthe next sentence. A device for transmission providing, in a knownfrequency band, m transmission channels each corresponding to a signalto be sent from the input of at least one transmitting circuit to theoutput of at least one receiving circuit, where m is an integer greaterthan or equal to 2, comprises:

-   -   an interconnection with n transmission conductors and a        reference conductor, n being an integer greater than or equal to        m, the interconnection being proportioned in such a way that the        interconnection may, taking into account the lumped impedances        seen by the interconnection and caused by the circuits connected        to the interconnection elsewhere than at the ends of the        interconnection, be modeled as a multiconductor transmission        line having uniform electrical characteristics over its length        for the known frequency band;    -   one or two termination circuits, each arranged at one end of the        interconnection and each having, in the known frequency band, an        impedance matrix approximating the characteristic impedance        matrix of the multiconductor transmission line, the termination        circuits, if there are several termination circuits, being each        arranged at a different end of the interconnection;    -   at least one of the transmitting circuits for combining the m        input signals, without using a transformer for this purpose,        according to linear combinations defined by a transition matrix        from modal electrical variables to natural electrical variables,        in order to obtain at the output of said one of the transmitting        circuits, output being connected to the n transmission        conductors, the generation of modal electrical variables, each        being proportional to a single signal among the input signals;        and    -   at least one of the receiving circuits, the input of which is        connected to the n transmission conductors for combining,        without using a transformer for this purpose, the signals        present on the transmission conductors, according to linear        combinations defined by the inverse of the transition matrix        from modal electrical variables to natural electrical variables,        so as to obtain at the output of said one of the receiving        circuits m output signals each corresponding to one of the        transmission channels, each output signal being proportional to        a single modal electrical variable among the modal electrical        variables.

Note that, as mentioned above, a device for implementing the method ofthe invention may:

-   -   either comprise only at one end of the interconnection, a        termination circuit with an impedance matrix near the        characteristic impedance matrix,    -   or comprise at both ends of the interconnection, a termination        circuit with an impedance matrix near the characteristic        impedance matrix.

In a device for implementing the method of the invention, it is possibleto obtain the generation of m modal electrical variables at the outputof a transmitting circuit.

In a device for implementing the method of the invention, it is possiblethat the number m of transmission channels between any one of thetransmitting circuits and any one of the receiving circuits is equal tothe number n of transmission conductors.

In the case where m=n, each mode has a corresponding transmissionchannel. Let X_(I) be the column-vector of the n input signals x_(I1), .. . , x_(In) of a transmitting circuit, and let X_(O) be thecolumn-vector of the n output signals x_(O1), . . . , x_(On) of areceiving circuit. These signals may be voltages or currents. Accordingto the invention, there is a proportionality between each modal voltageproduced by a transmitting circuit and the input signal of thecorresponding channel. Using a suitable numbering of the input signals,we may therefore write:V _(M)=diag_(n)(α₁, . . . , α_(n))X ₁   (13)where V_(M) is the vector of the modal voltages produced by thetransmitting circuit, and diag_(n)(α₁, . . . , α_(n)) is the diagonalmatrix of the non-zero proportionality coefficients α_(i). Thedimensions of each of these coefficients depend upon the dimensions ofthe input signals; if for instance these input signals are voltages, thecoefficients α_(i) will be dimensionless. Therefore, using the equation(6), we see that the transmitting circuit must produce, on eachconductor, at its point of connection to the interconnection, thenatural voltages of the vector V given by:V=S diag_(n)(α₁, . . . , α_(n))X _(I)   (14)

Moreover, given that, for each channel, a receiving circuit produces atits output, a signal practically proportional to the modal voltagecorresponding to the channel, we may, with a suitable numbering of theoutput signals, write that:X ₀=diag_(n)(β₁, . . . , β_(n))V _(M)   (15)where V_(M) is the vector of the modal voltages received by thereceiving circuit, and diag_(n)(β₁, . . . , β_(n)) is the diagonalmatrix of the non-zero proportionality coefficients β_(i). Thedimensions of these coefficients depend upon the dimensions of theoutput signals: if for instance the output signals are currents, β_(i)will have the dimensions of admittance. We see that the receivingcircuit must read the set of conductors, to obtain the modal voltages byapplying equation (6). Therefore, if, at the connection point of thereceiving circuit to the interconnection, the vector of the naturalvoltages is V, the output signals are given by:X _(o)=diag_(n)(β₁, . . . , β_(n))S ⁻¹ V   (16)

Given that, according to the invention, the waves propagate on theinterconnection as they would in a uniform multiconductor transmissionline, without significant reflection at the ends, it is possible, usingequations (13) and (15), to clarify how the transmission of signalstakes place. Between a transmitting circuit and a receiving circuitwhose connection points to the interconnection show a difference ofcurvilinear abscissa ΔL, for any integer i between 1 and n included, weobtain:x_(Oi)=α_(i)β_(i)e^(−γis 1) ^(|ΔL|)x_(Ii)   (17)

According to (14), this transmitting circuit must produce, for any oneof its output terminals i, the linear combination of the signals of theinput channels using the coefficients of the i-th line-vector of thematrix obtained by multiplying each column j of the matrix S by acoefficient α_(j). According to (16), the receiving circuit mentionedabove must produce, for any one of its output channels i, the linearcombination of the voltages at its input terminals using thecoefficients of the i-th line-vector of the matrix S⁻¹ multiplied by thecoefficient β_(i).

The person skilled in the art knows that such linear combinations, may,for instance, be implemented with operational amplifiers (in thefrequency bands where devices of this type may operate) and circuitelements of suitable impedance. Transformers can also be used to obtainsome linear combinations, but it is difficult and expensive to obtainall linear combinations using transformers. Moreover, transformers havea limited passband (for instance three decades of frequency), and theyblock DC. This is why, in the wording of the method of the invention,the use of transformers is excluded for obtaining linear combinations.

A device for implementing the method of the invention may be such thatthe electrical variables are either all voltages or all currents, andthe two formulations are in fact equivalent. Instead of equations (13)to (16), note that we could also have written, for a transmittingcircuitI _(M)=diag_(n)(α₁, . . . , α_(n))X _(I)   (18)I=T diag(α₁, . . . , α_(n))X _(I)   (19)where I_(M) is the vector of the modal currents produced by thetransmitting circuit, I is the vector of the corresponding naturalcurrents, and diag_(n)(α₁, . . . , α_(n)) is the diagonal matrix of thenon-zero proportionality coefficients α_(i), and, for the receivingcircuitX ₀=diag_(n)(β₁, . . . , β_(n))I _(M)   (20)X ₀=diag_(n)(β₁, . . . , β_(n))T ⁻¹ I   (21)where I_(M) is the vector of the modal currents received by thereceiving circuit, I is the vector of the corresponding naturalcurrents, and diag_(n)(β₁, . . . , β_(n)) is the diagonal matrix of thenon-zero proportionality coefficients β_(i). The proportionalitycoefficients appearing in equations (13) to (17) are obviously not thesame as the ones appearing in equations (18) to (21).

In a device for implementing the method of the invention, it is possiblethat the section of the interconnection in a plane orthogonal to thedirection of propagation does not change, except for a scale factor,over the greatest part of the length of the interconnection, in thevicinity of the transmission conductors.

A device for implementing the method of the invention may in particularbe such that n is greater than or equal to three.

A device for implementing the method of the invention may preferentiallybe such that the known frequency band contains frequencies between 100kHz and 100 GHz.

We have already mentioned that it is often possible, for instance atfrequencies greater than 100 kHz, to obtain real andfrequency-independent matrices Z_(C), S and T. In this case, it is clearfor the person skilled in the art that a termination circuit having, inthe known frequency band, an impedance matrix approximating thecharacteristic impedance matrix, could for instance be made of a networkof resistors, and the computations needed to proportion this network arenot difficult.

A device for implementing the method of the invention may be such thatthe termination circuit or the termination circuits are made of anetwork of resistors.

Termination circuits made of a network of resistors are however not atall a characteristic of the invention. By way of example, as mentionedin the Prior Art section, designers may, in order to limit the powerconsumed by a signal present at the terminals of termination circuits,choose to allow these terminals to be effective only in a relevantinterval of frequency, for instance by including suitable reactivecircuit elements in the termination circuits. Another example is thatterminations circuits could include active circuit elements.

In the case where real matrices Z_(C), S and T are considered, and wherethe chosen coefficients α_(i) and β_(i) of the equations (13) to (21)are real, it is also clear for the person skilled in the art that thelinear combinations specified in the transmitting circuits and in thereceiving circuits may be built using operational amplifiers (in thefrequency bands where devices of this type may operate) and resistors.However, at relatively high frequencies, unwanted phase differences inoperational amplifier circuits may become unavoidable, and may possiblyproduce non-real values for coefficients α_(i) and β_(i).

In the case where it may be useful to take losses into account whendetermining the matrices Z_(C), S and T, it is likely that thesematrices are not real and frequency-independent any more, and it becomesnecessary to synthesize the termination circuits and/or receivingcircuits and/or transmitting circuits, using methods well known to thepersons skilled in the art, such that the synthesized circuits includereactive circuit elements. Such a synthesis may for instance implementactive circuit elements.

According to the invention, it is specified that it must be possible tomodel the interconnection as a multiconductor transmission line havinguniform electrical characteristics over its length for the knownfrequency band, taking into account the lumped impedances seen by theinterconnection and caused by the circuits connected to it elsewherethan at its ends. In order to take these lumped impedances into accountby merely stating they are not present or that they may be ignored,these circuits must be such that they do not disturb the propagationalong the multiconductor transmission line. The person skilled in theart sees that this result can for instance be obtained by:

-   -   using transmitting circuits and/or receiving circuits connected        in series to the conductors of the interconnection, and showing        a low series impedance    -   using transmitting circuits and/or receiving circuits connected        in parallel to the conductors of the interconnection, and        showing a high parallel impedance.

A device for implementing the method of the invention may therefore besuch that the transmitting circuit(s) and the receiving circuit(s) areconnected in parallel to the interconnection, and such that theinterconnection sees a high impedance in the connections of thetransmitting circuit(s) and the receiving circuit(s). In this case, thedesigner may well consider that the transmitting circuit operates as acurrent source, and use equation (19). Alternatively, the designer mayuse the voltage angle and apply equation (14) considering that:

-   -   if the device for implementing the method of the invention is        such that only one end of the interconnection is connected to a        termination circuit with an impedance matrix near the        characteristic impedance matrix, then the other end is connected        to the transmitting circuit having a high impedance,        consequently the output of the transmitting circuit sees an        impedance matrix near Z_(C), and therefore        I=Z _(C) ⁻¹ S diag_(n)(α₁, . . . , α_(n))X _(I)   (22)    -   if the device for implementing the method of the invention is        such that both ends of the interconnection are connected to a        termination circuit with an impedance matrix near the        characteristic impedance matrix, then the output of the        transmitting circuit sees an impedance matrix near Z_(C)/2, and        therefore        I=2Z_(C) ⁻¹ S diag_(n)(α₁, . . . , α_(n))X _(I)   (23)

The designer shall of course keep in mind that diag_(n)(α₁, . . . ,α_(n)) does not mean the same thing in equation (19) and in equations(14), (22) and (23).

However, the connection of the transmitting circuits and/or thereceiving circuits in parallel with the interconnection is not at all acharacteristic of the invention. According to the invention, thetransmitting circuit(s) and/or the receiving circuit(s) may be connectedin series to the interconnection, in which case they must generally showa low series impedance to the interconnection, in order not to disturbthe propagation of waves along the interconnection. The seventhembodiment of a device of the invention, described below as an example,comprises a transmitting circuit connected in series to theinterconnection.

A device of the invention may be such that the termination circuits, thetransmitting circuit(s), and the receiving circuit(s) are without anypart in common to any two of them.

Conversely, a device of the invention may be such that the terminationcircuits, the transmitting circuit(s), and the receiving circuit(s) arenot without a part or parts common to any two of them. This possibilitywill be discussed below in the presentation of the fourth, fifth andsixth embodiments provided as examples.

According to the prior art, the person skilled in the art knows that thecrosstalk in an interconnection having parallel conductors is low at lowfrequencies, and that it strongly depends both on the frequency and onthe length of the interconnection. These properties usually limit themaximum length of the interconnection and its maximum frequency ofoperation. A device of the invention, in which we observe that thecrosstalk, that can be calculated, depends little on the frequency andthe length of the interconnection, does away with these limitations.

The prior art required, for obtaining very low crosstalk oninterconnections, that they have a complex three-dimensional structure,that they contain for instance a twisted pair for each channel, or ascreen for each channel. According to the invention, reduced crosstalkmay be obtained merely by creating an interconnection with parallelconductors, hence cutting down cost and size.

Finally, we note that according to the prior art, the desiredpropagation of a signal on a single conductor corresponds to thepropagation of several modes, at different propagation velocities,causing a modal dispersion well known to specialists. In the timedomain, this modal dispersion distorts the signals. According to theinvention, each signal is propagated using a single mode. There istherefore no modal dispersion, which increases the passband of theinterconnection and the maximum length it may have.

BRIEF DESCRIPTION OF THE DRAWINGS

Other advantages and characteristics will appear more clearly from thefollowing description of particular embodiments of the invention, givenby way of non-limiting examples, with reference to the accompanyingdrawings in which:

FIG. 1 shows an interconnection having 4 parallel transmissionconductors, which has already been discussed in the section dedicated tothe presentation of prior art;

FIG. 2 shows an interconnection linking a plurality of line transmittersand receivers, which has already been discussed in the section dedicatedto the presentation of prior art;

FIG. 3 shows a first embodiment of the invention;

FIG. 4 shows a second embodiment of the invention (best mode);

FIG. 5 shows a detail of a third embodiment of the invention;

FIG. 6 shows the symbols used in FIGS. 7 to 10;

FIG. 7 shows a fourth embodiment of the invention;

FIG. 8 shows a fifth embodiment of the invention;

FIG. 9 shows a sixth embodiment of the invention;

FIG. 10 shows a seventh embodiment of the invention.

DETAILED DESCRIPTION OF SOME EMBODIMENTS First Embodiment

As a first example of a device for implementing the method of theinvention, given by way of non-limiting example, we have represented inFIG. 3 a device of the invention comprising an interconnection (1)having four parallel transmission conductors and a reference conductor.The transmission conductors numbered 1, 2, 3 and 4 (this numbering isnot shown in FIG. 3) may be the conductors of a flat cable fitted with ascreen (or shielding), this screen being used as reference conductor. InFIG. 3, the two ends of the interconnection are each connected to atermination circuit (4) presenting an impedance matrix approximating thecharacteristic impedance matrix in a known frequency band. Thetransmitting circuit (5) receives at its input the signals of the fourchannels of the source (2), and its four output terminals are connectedto the conductors of the interconnection, this transmitting circuitproducing modal voltages on these conductors, each modal voltage beingproportional to the signal of a different channel. The receiving circuit(6) has its input terminals connected to the conductors of theinterconnection, this receiving circuit (6) producing four signals atits output terminals connected to the destination (3), each signal beingproportional to one of the modal voltages appearing on these conductors.Thus, the signals of the four channels of the source (2) are sent to thefour channels of the destination (3), without noticeable crosstalk.

Note that, in the device of FIG. 3, the receiving circuit (6) must besuch that its connection in parallel to the termination circuit (4) doesnot significantly alter the values of the impedance matrix connected tothe end of the line. The receiving circuit (6) must therefore present ahigh impedance to the interconnection (1), such that the interconnection(1) indeed sees at this end an impedance matrix nearing that of thetermination circuits (4).

Note that, in the device of FIG. 3, the transmitting circuit (5) can onthe contrary show any impedance to the interconnection (1), because noincident wave can reach the end of the interconnection (1) to which thetransmitting circuit (5) is connected. For that very reason, thetermination circuit (4) connected to the same end of the interconnectionas the transmitting circuit (5) could be removed, the advantage beingthat the transmitting circuit (5) would see twice the initial impedance,and that it would need to deliver only half the initial power to producea given voltage level at the receiving circuit (6).

Second Embodiment (Best Mode)

As a second example of a device for implementing the method of theinvention, given by way of non-limiting example and best mode ofcarrying out the invention, we have represented in FIG. 4 a device ofthe invention, comprising an interconnection (1) having four paralleltransmission conductors, and a reference conductor. The interconnectionis connected at each end to a termination circuit (4). Two transmittingcircuits (5) placed at two different abscissa z, receive at their inputsthe signals from the four channels of each of the two sources (2), thesetransmitting circuits (5) producing, when they are active, modalvoltages, each being proportional to the signal of one channel. Notethat this is a data bus architecture, and that the signals needed toobtain the active state of at most one transmitting circuit at a giventime are not shown in FIG. 4. The three receiving circuits (6) placed atthree different abscissa z, have their input terminals connected to theconductors of the interconnection, these receiving circuits (6)producing output signals being each proportional to a different modalvoltage, at their output terminals connected to the destinations (3).Thus, the signals of the four channels of a source (2), connected to anactive transmitting circuit (5), are sent to the four channels of thedestination (3) without noticeable crosstalk.

Note that, in the device of FIG. 4, the transmitting circuits (5) andthe receiving circuit (6), being connected in parallel to theinterconnection (1), must present a high impedance to theinterconnection (1), in order not to disturb the propagation of wavesalong the interconnection in a detrimental way, and in order not toproduce undesirable reflections at the ends of the interconnection (1).In the device of FIG. 4, both termination circuits are necessary becausewaves coming from the interconnection (1) may be incident on both ends.

Third Embodiment

As a third example of a device for implementing the method of theinvention, given by way of non-limiting example, we have considered theapplication of the method of the invention inside an integrated circuitin gallium arsenide technology, confining ourselves to discussing thecreation of the termination circuit(s). A paper by J. Chilo entitled“Modélisation et analyse temporelle d'un bus d'interconnexion entechnologie GaAs”, published in Annales des télécommunications, vol. 40,No. 3-4, Mars-Avril l985, provides the L and C matrices of such aninterconnection with 8 transmission conductors: $L = {\begin{pmatrix}{0,57} & {0,21} & {0,11} & {0,06} & {0,04} & {0,03} & {0,02} & {0,01} \\{0,21} & {0,57} & {0,21} & {0,11} & {0,06} & {0,04} & {0,03} & {0,02} \\{0,11} & {0,21} & {0,56} & {0,21} & {0,11} & {0,06} & {0,04} & {0,03} \\{0,06} & {0,11} & {0,21} & {0,56} & {0,21} & {0,11} & {0,06} & {0,04} \\{0,04} & {0,06} & {0,11} & {0,21} & {0,56} & {0,21} & {0,11} & {0,06} \\{0,03} & {0,04} & {0,06} & {0,11} & {0,21} & {0,56} & {0,21} & {0,11} \\{0,02} & {0,03} & {0,04} & {0,06} & {0,11} & {0,21} & {0,57} & {0,21} \\{0,01} & {0,02} & {0,03} & {0,04} & {0,06} & {0,11} & {0,21} & {0,57}\end{pmatrix}\mu\quad H\text{/}m}$ $C = {\begin{pmatrix}465 & {- 156} & {- 16} & {- 5} & {- 3} & {- 2} & {- 1} & {- 1} \\{- 156} & 523 & {- 150} & {- 14} & {- 4} & {- 2} & {- 1} & {- 1} \\{- 16} & {- 150} & 523 & {- 150} & {- 14} & {- 4} & {- 2} & {- 2} \\{- 5} & {- 14} & {- 150} & 523 & {- 150} & {- 14} & {- 4} & {- 3} \\{- 3} & {- 4} & {- 14} & {- 150} & 523 & {- 150} & {- 14} & {- 5} \\{- 2} & {- 2} & {- 4} & {- 14} & {- 150} & 523 & {- 150} & {- 16} \\{- 1} & {- 1} & {- 2} & {- 4} & {- 14} & {- 150} & 523 & {- 156} \\{- 1} & {- 1} & {- 2} & {- 3} & {- 5} & {- 16} & {- 156} & 465\end{pmatrix}{pF}\text{/}m}$

The reference conductor of this interconnection is made of twoconductive planes, which must be sufficiently interconnected, asexplained above. With this data, we can use equation (7) to calculatethe characteristic impedance matrix of the interconnection. We obtain:$Z_{C} = {\begin{pmatrix}{37,35} & {13,31} & {6,45} & {3,36} & {2,08} & {1,44} & {0,93} & {0,51} \\{13,31} & {37,07} & {13,19} & {6,38} & {3,32} & {2,04} & {1,40} & {0,93} \\{6,45} & {13,19} & {36,71} & {13,18} & {6,38} & {3,32} & {2,04} & {1,44} \\{3,36} & {6,38} & {13,18} & {36,71} & {13,18} & {6,38} & {3,32} & {2,08} \\{2,08} & {3,32} & {6,38} & {13,18} & {36,71} & {13,18} & {6,38} & {3,36} \\{1,44} & {2,04} & {3,32} & {6,38} & {13,18} & {36,71} & {13,19} & {6,45} \\{0,93} & {1,40} & {2,04} & {3,32} & {6,38} & {13,19} & {37,07} & {13,31} \\{0,51} & {0,93} & {1,44} & {2,08} & {3,36} & {6,45} & {13,31} & {37,35}\end{pmatrix}\Omega}$

Proportioning a network of resistors presenting an impedance matrixequal to Z_(C) may be achieved with methods well known to the personskilled in the art. The matrix Z_(C) being symmetrical, these exactmethods synthesize a network comprising at least n(n+1)/2 resistors,that is to say 36 resistors if n=8.

In fact, a termination circuit comprising far fewer resistors, providinga good approximation of Z_(C) can be defined easily. For instance, theschematic diagram of FIG. 5 illustrates such a termination circuit (4)made of 21 resistors. The input terminals (499) of this terminationcircuit are intended to be connected to the transmission conductors andits ground is intended to be connected to the reference conductor. Thevalues of the grounded resistors (401), (402), (403), (404), (405),(406), (407) and (408) range from 55 Ω to 89 Ω. The values of theresistors connected between adjacent transmission conductors, that is tosay the resistors (412), (423), (434), (445), (456), (467) and (478),range from 95 Ω to 100 Ω. The values of the resistors connected betweentransmission conductors having only a single other transmissionconductor between them, that is to say the resistors (413), (424),(435), (446), (457) and (468), range from 680 Ω to 770 Ω.

This possibility of reducing the number of parts of a terminationcircuit is related to the fact that, for this interconnection, thecouplings between distant conductors become quite weak. However, as isthe case for any approximation, it should be established if it isappropriate for a given performance objective.

Fourth Embodiment

Before going into the details of a fourth device of the invention, it isuseful to refer to FIG. 6, in which we have represented two symbols usedin the figures that follow after, namely the symbol of the voltagecontrolled voltage source (100) and the symbol of the voltage controlledcurrent source (200).

The voltage controlled voltage source (100) is an ideal circuit elementwell known to the person skilled in the art, implemented in the SPICEsimulation software of the University of California at Berkeley. Thisideal circuit element is characterized by its gain. The potentialdifference between the positive output terminal (103) and the negativeoutput terminal (104) is equal to the gain multiplied by the potentialdifference between the positive input terminal (101) and the negativeinput terminal (102).

The voltage controlled current source (200) is also an ideal circuitelement well known to the person skilled in the art, implemented in theSPICE simulation software. This ideal circuit element is characterizedby its transconductance. The current leaving its positive outputterminal (203) is equal to the current entering its negative outputterminal (204) and is equal to the transconductance multiplied by thepotential difference between its positive input terminal (201) and itsnegative input terminal (202).

The person skilled in the art knows how to create circuits having abehavior very similar to that of the voltage controlled voltage source(100) or the voltage controlled current source (200). There are manypossible schematic diagrams, depending mainly on the desired accuracyand on the operating frequency band. Note that the outputs of thevoltage controlled voltage source and of the voltage controlled currentsource are floating outputs. If an ideal circuit element is used withone of its outputs grounded, it is obviously not necessary to plan for acircuit providing the ideal circuit element function to have floatingoutputs. Otherwise, this characteristic may be obtained for instance byusing the concept of floating operational amplifier described in thepaper “Operational floating amplifier” by J. H. Huijsing, published inthe journal IEE Proceedings, Vol. 137, Pt. G, No. 2, April 1990.

Important aspects of the invention will appear more clearly from thefollowing description of a fourth embodiment, given by way ofnon-limiting example, and shown in the schematic diagram in FIG. 7. Thisdevice comprises a 30 cm long interconnection (1) having two paralleltransmission conductors and a reference conductor. For this particularinterconnection, the matrices L and C are $L = {\begin{pmatrix}{0,8629} & {0,3725} \\{0,3725} & {0,8629}\end{pmatrix}\mu\quad H\text{/}m}$ $C = {\begin{pmatrix}{46,762} & {{- 18},036} \\{{- 18},036} & {46,762}\end{pmatrix}{pF}\text{/}m}$and the losses are negligible. We may determine the matrices Z_(C), Sand T as explained above, and we obtain for instance:$Z_{C} = {\begin{pmatrix}{147,187} & {60,1923} \\{60,1923} & {147,187}\end{pmatrix}\Omega}$ $S = \begin{pmatrix}{1,0912} & {2,4616} \\{{- 1},0912} & {2,4616}\end{pmatrix}$ $T = \begin{pmatrix}{0,70711} & {0,70711} \\{{- 0},70711} & {0,70711}\end{pmatrix}$In the last two expressions, the matrices S and T are associated, withthe value of the arbitrary per-unit-length capacitance c_(K) defined byequation (5) equal to 10⁻¹⁰ F/m.

In FIG. 7, only one end of the interconnection (1) is connected to atermination circuit (4) made of three resistors (401), (402) and (403),the value of the resistors (401) and (402) being 207 Ω, and the value ofthe resistor (403) being 300 Ω, as these values produce an impedancematrix very close to Z_(C). The transmitting circuit (5) comprises twovoltage controlled current sources (511) and (512) and six resistors(521), (522), (523), (524), (525) and (526). This transmitting circuitreceives at its input the signal of the two channels of the source (2),represented by the voltage sources (21) and (22). The receiving circuit(6) comprises two voltage controlled voltage sources (611) and (612) andtwo resistors (621) and (622). These two resistors must not prevent theinterconnection from seeing a termination showing an impedance matrixapproximating the characteristic impedance matrix. As a consequence, thetwo resistors (621), (622) must each have a much greater value than thevalue of the resistor (403) and/or the receiving circuit must beconsidered as having a part in common with the termination circuit,which modifies the value of the resistor (403) in such a way that thisresistor in parallel with the resistors (621) and (622) in seriestogether provide the desired value of 300 Ω.

This schematic diagram and the proportioning of the circuit elements canbe directly inferred from the theory presented above. For instance, thevalues of the resistors of the transmitting circuit and of the receivingcircuit, the transconductance of the two voltage controlled currentsources and the gain of the two voltage controlled voltage sources, maybe inferred from the equations (16) and (22), upon choosing theproportionality coefficients α_(i) and β_(i) suited to the amplitudesthat the designer wishes on the interconnection.

With such proportioning, the transmitting circuit (5) would produce twomodal voltages, each being proportional to the signal produced by one ofthe voltage sources (21) and (22), and the receiving circuit (6) wouldproduce on the two output channels connected to the destination (3)represented by the resistors (31) and (32), two signals being eachproportional to a modal voltage. The signals of the two source (2)channels are sent to the two destination (3) channels, and a computationshows that there is no noticeable crosstalk. Of course, theproportioning of the transmitting circuit (5), the receiving circuit (6)and the termination circuit (4) does not depend on the length of theinterconnection.

The person skilled in the art knows how to build devices whose workingis fairly close to the schematic diagram of FIG. 7. For instance, adevice comprising a transmitting circuit and a receiving circuit,capable of operating in the frequency band of 100 kHz to 100 MHz, mayimplement fast operational amplifiers and current mirrors as activecircuit elements, and include no inductive circuit element. The absenceof an inductive circuit element, and in particular of a transformer,allows for an easy integration of such a circuit, for instance using asingle integrated circuit for the transmitting circuit (5) and a singleintegrated circuit for the receiving circuit (6) and the terminationcircuit (4).

It is interesting to note that this very example of interconnection wasdiscussed by C. R. Paul in his paper “Solution of the Transmission-LineEquation Under the Weak-Coupling Assumption”, published in the journalIEEE Transactions on Electromagnetic Compatibility, vol. 44, No. 3,August 2002, pages 413 to 423. Note that, in accordance with the priorart presented above, he calls “matched termination” a termination madeof just 135.8 Ω grounded resistors, and that with these resistors, hecomputes significant crosstalk shown in his FIG. 6. In this paper, heobserves that grounded resistors cannot be considered as truly matchedterminations, and he refers to his book cited above, in which heexplains in paragraph 5.2.6.1 that truly matched terminationsunavoidably produce high crosstalk. The theoretical contribution of theinvention is clearer now: it proves that, contrary to prior beliefs,with truly matched terminations such as the termination circuits of theinvention, provided that they are used in a suitable context, that is tosay with transmitting circuits and receiving circuits of the invention,we can practically eliminate crosstalk.

Fifth Embodiment

Other aspects of the invention may become more apparent from thefollowing description of a fifth embodiment, given by way ofnon-limiting example, and shown in the schematic diagram in FIG. 8. Thisdevice comprises an interconnection (1) having two transmissionconductors, identical to the one used in the fourth embodiment of theinvention.

In FIG. 8, the two ends of the interconnection (1) are each connected toa termination circuit (4) made of three resistors (401), (402) and(403), the value of the resistors (401) and (402) being 87 Ω and thevalue of the resistors (403) being 60,2 Ω, because these values indeedproduce an impedance matrix very close to Z_(C). Each of the twotransmitting circuits (5) has parts in common with a termination circuit(4) and comprises only the voltage controlled current sources (511) and(512) as its own circuit elements. These transmitting circuits (5)receive at their input the signals of the two channels of the twosources (2), each represented by the voltage sources (21) and (22).These transmitting circuits indeed produce modal voltages such that eachof them is proportional to the signal of one of the voltage sources (21)or (22). It is therefore essential that only one of the two sources (2)be active at any given time.

The two receiving circuits (6) have parts in common with each of thetermination circuits (4) and comprise only the voltage controlledvoltage sources (611) and (612) as their own circuit elements. Thesereceiving circuits indeed produce, on their two output channelsconnected to the destinations (3) each represented by the resistors (31)and (32), two signals being each proportional to a modal voltage. Thesignals of the two channels of an active source (2) are sent to the twochannels of the two destinations (3), and a computation shows that thereis no noticeable crosstalk.

It is interesting to compare this embodiment to the previous one,because we can see that devices of the invention comprising the sameinterconnection may have quite different schematic diagrams. Inparticular, note that, in the device in FIG. 7, the termination circuit(4), the transmitting circuit (5) and the receiving circuit (6) are allseparate, whereas, in the device in FIG. 8, the termination circuits(4), the transmitting circuits (5) and the receiving circuits (6) haveparts in common, which allows for a particularly small number of circuitelements. The person skilled in the art will also note that thesimplicity of schematic diagram in FIG. 8 is related to the fact thatthe interconnection is balanced, as observed when examining the matricesL and C.

Sixth Embodiment

Other aspects of the invention may appear more clearly from thefollowing description of a sixth embodiment, given by way ofnon-limiting example, and shown in the schematic diagram in FIG. 9. Thisdevice comprises an interconnection (1) having three paralleltransmission conductors and one reference conductor, 40 cm in length.For this particular interconnection, the matrices L and C are$L = {\begin{pmatrix}{0,3139} & {0,0675} & {0,0222} \\{0,0675} & {0,3193} & {0,0675} \\{0,0222} & {0,0675} & {0,3139}\end{pmatrix}\mu\quad H\text{/}m}$ $C = {\begin{pmatrix}{130,3} & {{- 16},2} & {{- 0},8} \\{{- 16},2} & {133,7} & {{- 16},2} \\{{- 0},8} & {{- 16},2} & {130,3}\end{pmatrix}{pF}\text{/}m}$and the losses will be assumed negligible. An interconnection with theseparameters has been discussed by J. G. Nickel, D. Trainor and J. E.Schutt-Ainé in their paper “Frequency-Domain-Coupled Microstrip-LineNormal-Mode Parameter Extraction From S-Parameters”, published in thejournal IEEE Transactions on Electromagnetic Compatibility, vol. 43, No.4, November 2001, pages 495 to 503. We may determine the matrices Z_(C),S and T as explained above, and we obtain for instance:$Z_{C} = {\begin{pmatrix}{49,41} & {8,35} & {2,24} \\{8,35} & {49,53} & {8,35} \\{2,24} & {8,35} & {49,41}\end{pmatrix}\Omega}$ $S = \begin{pmatrix}{0,3101} & {{- 0},5394} & {{- 0},4793} \\{{- 0},4755} & 0 & {{- 0},6232} \\{0,3101} & {0,5394} & {{- 0},4793}\end{pmatrix}$ $T = \begin{pmatrix}{0,4786} & {{- 0},7071} & {0,5198} \\{{- 0},7361} & 0 & {0,6780} \\{0,4786} & {0,7071} & {0,5198}\end{pmatrix}$In the last two expressions, the matrices S and T are associated, withthe value of the arbitrary per-unit-length capacitance c_(K) defined byequation (5) equal to 10⁻¹⁰ F/m.

In FIG. 9, only one end of the interconnection (1) is connected to atermination circuit (4) made of six resistors (401), (402), (403),(404), (405) and (406), the value of the resistors (401) and (403) being58,7 Ω, the value of the resistor (402) being 69,2 Ω, the value of theresistors (404) and (405) being 289,5 Ω and the value of the resistor(406) being 2781 Ω, because these values produce an impedance matrixvery close to Z_(C). The transmitting circuit (5) comprises threevoltage controlled voltage sources (511), (512) and (513) and tenresistors (521), (522), (523), (524), (525), (526), (527), (528), (529)and (530). This transmitting circuit receives a input the signal of thethree channels of the source (2), represented by the voltage sources(21), (22) and (23). The receiving circuit (6) comprises three voltagecontrolled voltage sources (611), (612) and (613) and seven resistors(621), (622), (623), (624), (625), (626) and (627). These sevenresistors must not prevent the interconnection from seeing a terminationpresenting an impedance matrix near the characteristic impedance matrix.These resistors must therefore have sufficiently large values and/or thereceiving circuit must be considered as having a part in common with thetermination circuit, which would modify the values defined above for thesix resistors (401), (402), (403), (404), (405) and (406) in such a waythat the interconnection (1) indeed sees its end connected to a networkhaving an impedance matrix approximating its characteristic impedancematrix.

This schematic diagram and the proportioning of the circuit elements canbe directly inferred from the theory presented above. For instance, thevalues of the resistors of the transmitting circuit and of the receivingcircuit, and the gain of the six voltage controlled voltage sources,maybe inferred from the equations (14) and (16), upon choosing theproportionality coefficients α_(i) and β_(i) suited to the amplitudesthat the designer wishes to obtain on the interconnection.

With such proportioning, the transmitting circuit (5) would producethree modal voltages, each being proportional to the signal on one ofthe input channels, and the receiving circuit (6) would produce on thethree output channels connected to the destination (3) represented bythe resistors (31), (32) and (33), three signals being each proportionalto a different modal voltage. The signals of the three channels of thesource (2) are sent to the three channels of the destination (3), and acomputation shows that there is no noticeable crosstalk

Seventh Embodiment

Other aspects of the invention may appear more clearly from thefollowing description of a seventh embodiment, given by way ofnon-limiting example, and shown on the schematic diagram in FIG. 10.This device comprises an interconnection (1) having four conductors,identical to the one used in the sixth embodiment.

As in the second embodiment represented in FIG. 4, we can see in FIG. 10a transmitting circuit (5) that is not at one end of theinterconnection. As explained above, this situation requires the use ofa termination circuit (4) at both ends of the interconnection. We alsonote that, contrary to the other embodiments presented above, in FIG. 10the transmitting circuit (5) is connected in series with the conductorsof the interconnection (1).

In FIG. 10, the two ends of the interconnection (1) are each connectedto a termination circuit (4) made of six resistors (401), (402), (403),(404), (405) and (406), having the same values as in the sixthembodiment. The transmitting circuit (5) comprises three voltagecontrolled voltage sources (511), (512) and (513) and ten resistors(521), (522), (523), (524), (525), (526), (527), (528), (529) and (530).This transmitting circuit receives at its input the signal of the threechannels of the source (2), represented by the voltage sources (21),(22) and (23). The receiving circuit (6) comprises three voltagecontrolled voltage sources (611), (612) and (613) and seven resistors(621), (622), (623), (624), (625), (626) and (627). As explained for thesixth embodiment, these seven resistors must not prevent theinterconnection from seeing a termination showing an impedance matrixnear the characteristic impedance matrix.

This schematic diagram and the proportioning of the circuit elements canbe directly inferred from the theory presented above. The values of theresistors of the transmitting circuit and of the receiving circuit, andthe gain of the six voltage controlled voltage sources, may be the sameas for the sixth embodiment. However, the person skilled in the artnotes that, if we want the sixth and the seventh embodiment to deliverthe same amplitude to the interconnection (1) and to the destination (3)for given signals coming from the three source (2) channels, the gainsof the three voltage controlled voltage sources (511), (512) and (513)of the transmitting circuit (5) must be, in the circuit of FIG. 10,twice as large as the corresponding gains used in the circuit of FIG. 9.

Here again, it can be proved, for instance with a simulation, that withsuch proportioning, the signals of the three source (2) channels aresent to the three destination (3) channels, without noticeablecrosstalk.

INDICATIONS ON INDUSTRIAL APPLICATIONS

According to the invention, it is possible to built into one or severalitems intended to be interconnected, for instance integrated circuits, asaid transmitting circuit and/or a said receiving circuit, to be usedfor interconnections having predetermined characteristics, for instancea required drawing of the section of the interconnection orthogonal tothe direction of propagation, for an implementation on an external layerof a printed circuit board using a substrate of epoxy-bonded fiberglassof specified permittivity. If such an item is used, we could obtain adevice for implementing the method of the invention, wherein one or moreitems to be interconnected contain a transmitting circuit and/or areceiving circuit, intended for interconnections with predeterminedcharacteristics. The designer integrating such items need only create aninterconnection of any length and of said predetermined characteristics,and the termination circuits, in order to obtain a device of theinvention. It is clear that this approach would be interesting forinstance for items intended to be connected to a data bus, e.g.microprocessors or memories, or for instance for circuit boards intendedto be connected to a backplane comprising the conductors of theinterconnection.

According to the invention, it is possible to built into one or severalitems intended to be interconnected, for instance integrated circuits, asaid transmitting circuit and/or a said receiving circuit, designed forinterconnections having predetermined characteristics, and also to buildinto these items a termination circuit designed for interconnectionshaving the same said predetermined characteristics. The designerintegrating such items need only create an interconnections having thepredetermined characteristics, to obtain a device of the invention.

According to the invention, with a reference conductor and amulticonductor transmission line having n transmission conductors, mtransmission channels are created, which allow the transmission of msignals. The number m is less than or equal to a n, but for a giveninteger n, it is always possible to design a device of the inventionsuch that m=n. For a specified number m of channels, it would seemunwise to choose a value of n that is greater than m. But we now seethat this circumstance is likely to occur if we wish to use a standardinterconnection having a fixed number of transmission conductors.

The invention is particularly suitable for cases where the transmissionchannels are used to send digital signals. In fact, in this case, aresidual crosstalk coupling factor is acceptable, but the bandwidth totake into account is often very wide. According to the invention, thisresult is easily obtained in an inexpensive way, because it iscompatible with the use of resistors of average accuracy.

As shown with the second embodiment, the invention is appropriate for animplementation wherein the interconnection is operated as a data bus.

The invention is particularly suitable for its implementation withmicrostrip structures and stripline structures, for instance on printedcircuit boards. It is particularly beneficial to printed circuit boardscomprising a wide band analog circuitry or fast digital circuits. Itsimplementation on printed circuit boards would for instance freedesigners of digital circuits from the limitations as to the length oftraces which they up to now had to observe.

The invention is therefore applicable when manufacturing computers,which include a large number of long interconnections for very fastsignals.

The invention is also particularly suitable for reducing crosstalk inflat multiconductor cables and inside integrated circuits.

1. A method for transmitting through an interconnection with ntransmission conductors and a reference conductor, n being an integergreater than or equal to 2, the method providing, in a known frequencyband, m transmission channels each corresponding to a signal to be sentfrom the input of at least one transmitting circuit to the output of atleast one receiving circuit, where m is an integer greater than or equalto 2 and less than or equal to n, the method comprising the steps of:modeling the interconnection, taking into account the lumped impedancesseen by the interconnection and caused by the circuits connected to theinterconnection elsewhere than at the ends of the interconnection, as amulticonductor transmission line having uniform electricalcharacteristics over its length for the known frequency band;determining, for the multiconductor transmission line and the knownfrequency band, the characteristic impedance matrix and a transitionmatrix from modal electrical variables to natural electrical variables;placing at at least one end of the interconnection, a terminationcircuit having an impedance matrix approximating the characteristicimpedance matrix; combining the m input signals in one of thetransmitting circuits without using a transformer for this purpose,according to linear combinations defined by the transition matrix frommodal electrical variables to natural electrical variables, so as toobtain at the output of said one of the transmitting circuits, outputbeing connected to the n transmission conductors, the generation ofmodal electrical variables, each being proportional to a single signalamong the input signals; and combining in one of the receiving circuits,the input of which is connected to the n transmission conductors,without using a transformer for this purpose, the signals present on thetransmission conductors, according to linear combinations defined by theinverse of the transition matrix from modal electrical variables tonatural electrical variables, so as to obtain at the output of said oneof the receiving circuits m output signals each corresponding to one ofthe transmission channels, each output signal being proportional to asingle modal electrical variable among the modal electrical variables.2. The method of claim 1, wherein the generation of m modal electricalvariables is obtained at the output of a transmitting circuit.
 3. Themethod of claim 1, wherein the number m of transmission channels betweenany one of the transmitting circuits and any one of the receivingcircuits is equal to the number n of transmission conductors.
 4. Themethod of claim 1, wherein the electrical variables are either allvoltages or all currents.
 5. The method of claim 1, wherein the sectionof the interconnection in a plane orthogonal to the direction ofpropagation does not change, except for a scale factor, over thegreatest part of the length of the interconnection, in the vicinity ofthe transmission conductors.
 6. The method of claim 1, wherein n isgreater than or equal to three.
 7. The method of claim 1, wherein theknown frequency band contains frequencies ranging from 100 kHz to 100GHz.
 8. A device for proportioning the circuits used in a method fortransmitting through an interconnection with n transmission conductorsand a reference conductor, n being an integer greater than or equal to2, the method providing, in a known frequency band, m transmissionchannels each corresponding to a signal to be sent from the input of atleast one transmitting circuit to the output of at least one receivingcircuit, where m is an integer greater than or equal to 2 and less thanor equal to n, the device comprising: means for modeling theinterconnection, taking into account the lumped impedances seen by theinterconnection and caused by the circuits connected to theinterconnection elsewhere than at the ends of the interconnection, as amulticonductor transmission line having uniform electricalcharacteristics over its length for the known frequency band; means fordetermining, for the multiconductor transmission line and the knownfrequency band, the characteristic impedance matrix and a transitionmatrix from modal electrical variables to natural electrical variables;means for proportioning a termination circuit having an impedance matrixapproximating the characteristic impedance matrix; means forproportioning one of the transmitting circuits which combines the minput signals, without using a transformer for this purpose, accordingto linear combinations defined by the transition matrix from modalelectrical variables to natural electrical variables, so as to obtain atthe output of said one of the transmitting circuits, output beingconnected to the n transmission conductors, the generation of modalelectrical variables, each being proportional to a single signal amongthe input signals; and means for proportioning one of the receivingcircuits, the input of which is connected to the n transmissionconductors, which combines, without using a transformer for thispurpose, the signals present on the transmission conductors according tolinear combinations defined by the inverse of the transition matrix frommodal electrical variables to natural electrical variables, so as toobtain at the output of said one of the receiving circuits m outputsignals each corresponding to one of the transmission channels, eachoutput signal being proportional to a single modal electrical variableamong the modal electrical variables.
 9. The device of claim 8 whereinthe means for modeling the interconnection comprise means for measuringand/or for computing the real electrical characteristics of theinterconnection, based on the relative layout of the transmissionconductors and the reference conductor, and on the characteristics ofthe dielectrics surrounding them.
 10. The device of claim 8, wherein themeans for modeling the interconnection comprise: means for calculatingone or more error coefficients for variance between the actualelectrical characteristics of the interconnection and the desiredcharacteristics, for the known frequency band; and means for optimizingthe relative position of the transmission conductors by minimizing theerror coefficients or coefficients.
 11. A device for transmissionproviding, in a known frequency band, m transmission channels eachcorresponding to a signal to be sent from the input of at least onetransmitting circuit to the output of at least one receiving circuit,where m is an integer greater than or equal to 2, the device comprising:an interconnection with n transmission conductors and a referenceconductor, n being an integer greater than or equal to m, theinterconnection being proportioned in such a way that theinterconnection may, taking into account the lumped impedances seen bythe interconnection and caused by the circuits connected to theinterconnection elsewhere than at the ends of the interconnection, bemodeled as a multiconductor transmission line having uniform electricalcharacteristics over its length for the known frequency band; one or twotermination circuits, each arranged at one end of the interconnectionand each having, in the known frequency band, an impedance matrixapproximating the characteristic impedance matrix of the multiconductortransmission line, the termination circuits, if there are severaltermination circuits, being each arranged at a different end of theinterconnection; at least one of the transmitting circuits for combiningthe m input signals, without using a transformer for this purpose,according to linear combinations defined by a transition matrix frommodal electrical variables to natural electrical variables, in order toobtain at the output of said one of the transmitting circuits, outputbeing connected to the n transmission conductors, the generation ofmodal electrical variables, each being proportional to a single signalamong the input signals; and at least one of the receiving circuits, theinput of which is connected to the n transmission conductors forcombining, without using a transformer for this purpose, the signalspresent on the transmission conductors, according to linear combinationsdefined by the inverse of the transition matrix from modal electricalvariables to natural electrical; variables, so as to obtain at theoutput of said one of the receiving circuits m output signals eachcorresponding to one of the transmission channels, each output signalbeing proportional to a single modal electrical variable among the modalelectrical variables.
 12. The device of claim 11 wherein the generationof m modal electrical variables is obtained at the output of atransmitting circuit.
 13. The device of claim 11, wherein the number mof transmission channels between any one of the transmitting circuitsand any one of the receiving circuits is equal to the number n oftransmission conductors.
 14. The device of claim 11, wherein the sectionof the interconnection in a plane orthogonal to the direction ofpropagation does not change, except for a scale factor, over thegreatest part of the length of the interconnection, in the vicinity ofthe transmission conductors.
 15. The device of claim 11, wherein n isgreater than or equal to three.
 16. The device of claim 11, wherein thetermination circuit(s) is(are) made of a network of resistors.
 17. Thedevice of claim 11, wherein the transmitting circuit(s) end thereceiving circuit(s) are connected in parallel to the interconnection,and wherein the interconnection sees a high impedance in the connectionsof the transmitting circuit(s) and the receiving circuit(s).
 18. Thedevice of claim 11, wherein one or more items to be interconnectedcontains a transmitting circuit and/or a receiving circuit, intended forinterconnections with predetermined characteristics.
 19. The device ofclaim 11, wherein the transmission channels are used to send digitalsignals.
 20. The device of claim 19, wherein the interconnection isoperated as a data bus.